Switching mode power amplifiers, such as Class-D and Class-E amplifiers, are widely used nowadays due to their high efficiency. However, as highly non-linear devices, switching mode power amplifiers also generate quite a high amount of energy at harmonic frequencies. Especially when using amplifiers with a single-ended output, the even-order harmonic tones are not cancelled out, as opposed to the amplifiers with an ideally differential output. The unwanted tones at the harmonic frequencies then violate the out-of-band spurious emission limits as defined by the ETSI or FCC standard regulation bodies.
Electronic amplifiers are typically classified into classes, e.g. Class-A, B, C, D, E, or even F. They can be grouped into two categories: linear amplifiers like Class-A, B, C and switching mode amplifiers like Class-D, E, F. The active element in a linear amplifier operates in the linear region, which means that the output signal is proportional to the input signal. A sinusoidal input signal results in a sinusoidal output with a larger amplitude. Switching mode amplifiers adopt a different operation principle. Normally, their input signal is a square wave to overdrive the active element. In this case the active element works like a switch, such that it is either completely on for conducting current or completely off. The output signal is then no longer proportional to the input signal. Notwithstanding the difference in operation principle, a conduction angle can be defined for these two categories in almost the same way. The conduction angle of an amplifier depends on the proportion of time during which the active element conducts current. For linear amplifiers the active element operates in the linear region when it is conducting and the output is proportional to the input. For switching amplifiers the active element works like a switch when conducting and the output is not proportional to the input. In the description below the conduction angle is referred to as 2α, so α represents half of the conduction angle (i.e. α ranges from 0 to π). If Class-A, the active element is always conducting, the conduction angle is 2π, and thus α equals π. If Class-B, the active element is conducting for only half of each cycle of the input signal, the conduction angle is π, and consequently α equals π/2. For switching mode amplifiers, the input signal is a square wave and the conduction angle is also equivalent to the duty cycle of the input square wave. If the duty cycle is 50%, the input voltage is high during half the period of each cycle and the active element conducts, so the conduction angle is π, and α equals to π/2.
The conduction angle of a switching mode amplifier is closely related to its harmonics emission performance. For an ideal switching mode amplifier with a broadband resistive load as shown in FIG. 1, the relationship between conduction angle and output power is illustrated in FIG. 2. As explained above, the conduction angle 2α means the period during which the switch is closed. Hence, α equals π/2, meaning that the switch is closed half of the time during the period of the input signal.
It is clear that the 2nd harmonic output power is zero when α is π/2, and the 3rd harmonic power becomes zero if α is π/3 or 2π/3. So if the conduction angle of the amplifier is well controlled, certain harmonic tones can be minimized. However, the real conduction angle of a practical amplifier is difficult to measure. For example, an amplifier for a WiFi system needs to work at 2.4 GHz. The period of a 2.4 GHz signal is 417 picoseconds. A conduction angle of π means the conduction time of the active element in the amplifier is only 208.5 picoseconds in each cycle. Directly measuring such a short time duration is extremely challenging.
One way to solve the harmonics emission problem would be to add an extra band-pass filter at the output of the amplifier to filter out any unwanted harmonics. However, such filter has to operate at radio frequencies (RF). High-performance RF band-pass filters are normally made of quartz or ceramics, which makes them difficult to be integrated with a power amplifier manufactured by modern semiconductor technology. Additionally, it introduces extra power loss. Another possible way would be to use a differential output topology to cancel out the even-order harmonics. However, due to poor matching of the two differential branches, the suppression effect is limited.
U.S. Pat. No. 5,712,593 aims at maximizing efficiency while maintaining an allowable distortion level over the entire dynamic range of the power amplifiers. FIG. 3 shows the block diagram of the disclosed circuit. Sampler 14 samples the output signal from amplifier 12 and provides the sampled signal to filter 16. Filter 16 filters out the desired signal and only passes through the undesired signals (harmonics or spurs) to detector 18. Detector 18 converts the filtered output signal to a DC signal proportional to the power level of the filtered output signal. Bias control portion 20 compares the DC signal provided by power detector 18 to a reference signal to produce a control signal. The control signal controls the saturation characteristics of power amplifier portion 12. Such a control loop continually re-biases the power amplifier for maximum efficiency for a desired level of distortion.
The scheme of FIG. 3 can however only be used for linear amplifiers. The active device in power amplifier 12 must be in its linear region, so that the bias condition can be controlled by the feedback loop to change its linearity. A band reject filter 16 is needed to capture the non-linearity of the output signal by selecting the unwanted signals to be suppressed. It selects the unwanted signals that shall be suppressed, and its frequency response determines the cancellation performance of the system. After this, detector 18 is needed to convert the RF signal from filter 16 to a readable DC signal. The presence of band reject filter 16 and detector 18 limits the flexibility of the proposed solution. First, the harmonic cancellation of the systems depends on the frequency response of the BPF, as it determines how well the unwanted signal is being filtered. Secondly, the value of the reference voltage depends on (i) the required distortion level as defined by the user and (ii) on the voltage transfer function (VTC) of detector 18, which makes the selection of the reference voltage technology dependent. Thus, selecting a proper reference voltage is not straightforward and requires complex calculation.
Hence, there is a need for a solution where the conduction angle can be controlled without undue burden, so that the influence a harmonic component can be reduced by adjusting the conduction angle.